1. Field of the Invention
Embodiments of the invention relate to radio frequency (RF) transceivers, and particularly to transceiver architectures employing multiple stages for modulation or demodulation of RF signals.
2. Related Technology
Transceiver circuits are used in devices that receive or transmit information using radio frequency modulation. Examples of devices that use transceiver circuits include wireless LAN interfaces, cell phones, personal digital assistants, GPS receivers and other devices having RF communication features.
Transceiver circuit architectures may be categorized by the manner in which they modulate and demodulate signals. FIG. 1 shows an example of a direct conversion receiver circuit that converts an RF signal directly to a baseband signal. In this circuit an RF signal received by an antenna 10 is filtered in a surface acoustic wave (SAW) filter 12 having a band pass profile for selecting a frequency band of interest. The filtered signal is provided to a balun 14 that converts the signal from a single ended signal to a differential signal. The differential signal is provided as input to a low noise amplifier 16 having a programmable gain. The low noise amplifier 16 provides the differential RF signals to respective mixers 18, 20 where they are down-converted to differential in-phase (I) and quadrature (Q) baseband signals by mixing with differential in-phase and quadrature signals generated by an RF local oscillator (LO) 22. The baseband signals are output from the mixers 18, 20 to respective low pass filters 24, 26 and then to baseband amplifiers 28, 30. A direct conversion transmitter circuit is shown in FIG. 2. In the transmitter circuit, differential in-phase and quadrature baseband signals are filtered by respective low pass filters 32, 34 and provided to mixers 36, 38 where they are up-converted to RF by mixing with differential in-phase and quadrature signals generated by an RF local oscillator 40. The outputs of the mixers 36, 38 are summed and provided to a balun 42 where they are converted from differential signals to a single ended signal. The output of the balun 42 is provided to a power amplifier 44 that drives an antenna 10.
The transmitter and receiver architectures of FIGS. 1 and 2 are referred to as direct conversion architectures because a single mixing stage is used to convert signals between RF and baseband. For purposes of channel selection in the receiver, the RF local oscillator frequency is varied so as to down-convert the frequency of the selected channel to baseband. Spurious tones from neighboring channels are eliminated from the baseband signal by the low pass filters 24, 26. Similarly, in the transmitter the RF local oscillator frequency is varied so as to up-convert the baseband signal to the desired RF channel frequency.
The direct conversion method is sometimes preferred for its relatively simple architecture, however it also has a number of drawbacks. For example, in order to provide high accuracy in the mixers, it is necessary for the mixers to receive differential inputs. This requires the use of a balun, which in turn requires the use of multiple inductors that increase the total size of the circuit. Also, because there are few amplifier stages, this architecture also requires the use of very low noise amplifiers, which necessitates large amplifier size and power consumption.
Other drawbacks of direct conversion architectures are specific to the local oscillator circuit. In the direct conversion architecture, the local oscillator circuit must produce exact frequencies for down-converting or up-converting specific RF channels. For example, in wireless LAN devices, a direct conversion local oscillator for the 802.11 b/g standards must produce frequencies that are adjustable in increments of 5 MHz from 2412 MHz to 2477 MHz, and also 2484 MHz. For the 802.11j standard, the local oscillator must produce frequencies that are adjustable in increments of 20 MHz from 4920 MHz to 4980 MHz, from 5040 MHz to 5080 MHz, and from 5170 MHz to 5230 MHz. For the 802.11a standard, the local oscillator must produce frequencies that are adjustable in increments of 20 MHz from 5180 MHz to 5320 MHz, and from 5745 MHz to 5845 MHz. The need for a local oscillator that is adjustable in this manner is a significant restraint on device performance because it requires the use of a local oscillator that generates significant phase noise.
FIG. 3 shows an example of a conventional phase locked loop circuit that is typically used as a local oscillator in circuits such as direct conversion transceivers. The primary frequency generating element of this circuit is a voltage controlled oscillator 50 that produces an output frequency corresponding to a charge stored in a low pass filter 52. The charge in the low pass filter 52 is controlled by a charge pump 54 that drives current into or out of the low pass filter 52 in response to a control signal provided by a phase frequency detector 56. The phase frequency detector 56 produces the control signal based on comparison of the frequency of the voltage controlled oscillator 50 to a reference frequency. The reference frequency is typically generated by dividing the signal from a reference frequency generator 58 such as a crystal oscillator using a frequency divider 60. The signal from the voltage controlled oscillator 50 is also typically divided by a frequency divider 62 so that the frequencies compared by the phase frequency detector 56 are approximately the same. The frequency dividers are a significant source of phase noise in this circuit and the phase noise increases as the divisor number of the frequency divider increases. In the case of the 802.11 a/j and 802.11 b/g standards, very large divisor numbers are required to produce the local oscillator frequencies that are needed for the channels of those standards. Consider an example in which a 40 MHz crystal oscillator is used as a reference frequency generator and the circuit must generate a frequency of 2417 MHz. In order to produce frequencies that can be compared by the phase frequency detector 56, the 40 MHz reference frequency and the 2417 MHz output frequency must each be divided down to a common factor of 1 MHz. This requires the frequency divider 62 that divides the VCO output frequency to utilize a divisor of 2417. This divisor is very large and results in the production of significant phase noise. Divisors of similar magnitudes must be used for the other channels of the 802.11 a/j and 802.11 b/g standards.
In addition to high phase noise, the high frequency local oscillators used in direct conversion architectures typically suffer from high flicker noise, significant skew between in-phase and quadrature signals, and significant DC offsets between the differential baseband signals. The need to produce in-phase and quadrature signals at higher frequencies also requires these local oscillators to utilize on-chip inductors that consume significant area.
An alternative to direct conversion transceiver architectures is the multistage or superheterodyne transceiver architecture. FIG. 4 shows an example of a conventional superheterodyne receiver circuit. In this circuit, a signal is received at an antenna 10 and is filtered by a SAW filter 12. The filtered signal is provided by the SAW filter 12 to a single ended programmable gain low noise amplifier 70. The amplifier 70 provides the RF signal to an RF mixer 72 where it is down-converted to a fixed intermediate frequency (IF) value by mixing with an RF local oscillator signal from an RF local oscillator 74. The intermediate frequency signal is output to a SAW filter 76 having a band select profile corresponding to the predetermined intermediate frequency. The filtered signal is provided to respective IF mixers 78, 80 where it is down-converted to differential in-phase and quadrature baseband signals by mixing with in-phase and quadrature signals from an intermediate frequency local oscillator 82. The differential baseband signals are filtered by low pass filters 84, 86 and amplified by programmable gain baseband amplifiers 88, 90. A corresponding superheterodyne transmitter architecture uses a fixed frequency local oscillator and mixers to up-convert differential in-phase and quadrature baseband signals to a fixed intermediate frequency, and then uses a variable frequency RF local oscillator and RF mixer to up-convert the intermediate frequency signal to an RF signal at the frequency of a selected channel.
During operation, the superheterodyne receiver down-converts the received RF signal to a fixed intermediate frequency and then performs a second down-conversion of the intermediate frequency to baseband. For purposes of channel selection, the RF local oscillator frequency is varied so as to down-convert the frequency of the selected channel to the fixed intermediate frequency, and spurious tones from adjacent channels are removed by the SAW filter 76.
The superheterodyne transceiver architecture has several advantages over the direct conversion architecture. High gain in the front end amplifier 70 and in the RF mixer 72 relaxes the noise requirements for the baseband amplifiers 88, 90 and therefore reduces the size and power requirements of the baseband amplifiers. Also, second order distortion in this circuit is less crucial and so a single ended amplifier topology may be used at the front end, resulting in a lower noise figure for the circuit as a whole. With regard to the local oscillators, the ability to use a single ended topology in the first down-conversion stage eliminates the need for the inductors required by the direct conversion mixer as well as the inductors required for a front-end balun. Also, the generation of in-phase and quadrature signals is much more accurate at intermediate frequencies than at direct conversion frequencies, and so the intermediate frequency local oscillator produces more accurate signals and results in lower DC offsets in the baseband signals.
Despite these advantages, the superheterodyne transceiver architecture also has several drawbacks. The need for a SAW filter 76 between the RF and IF mixing stages is a disadvantage because SAW filters are discrete components that are expensive and relatively large. Further, in order to reduce the introduction of noise from image channels, the RF mixer must be implemented as an image reject mixer, which consumes a relatively large amount of space and power. In addition, the need to down-convert the various channels within the RF band to a fixed intermediate frequency requires the RF local oscillator to be adjustable among values dictated by the channel frequencies and the intermediate frequency, leading to phase noise problems similar to those described in regard to direct conversion architectures.
One alternative to the superheterodyne architecture of FIG. 4 is sometimes referred to as a “sliding IF” architecture. The sliding IF architecture differs from the architecture of FIG. 4 in that the IF local oscillator is provided by dividing the RF local oscillator frequency by a fixed number. This causes the intermediate frequency to vary with changes in the RF local oscillator. While this architecture simplifies the design of the oscillators, it still requires the RF oscillator to switch among frequencies that require the use of large divisor numbers in the frequency synthesizer, and therefore phase noise continues to be a substantial problem.
Therefore both the direct conversion and superheterodyne architectures have shortcomings that limit device performance, size and expense.